1. Field of the Invention
The present invention relates generally to digital quadrature modulators, and more particularly, to digital quadrature modulators used as MODEMs for digital communication equipments such as a land mobile radio telephone, a portable radio telephone and a cordless telephone.
2. Description of the Background Art
A conventional digital communication apparatus modulates a carrier signal in response to a digital information signal (baseband signal) to transmit the information signal in order to achieve efficient transmission.
Such modulation systems include an amplitude modulation system wherein an amplitude of a carrier signal is changed in response to a digital baseband signal (a modulating wave signal), a frequency modulating system wherein a frequency of a carrier is deviated in response to a modulating wave signal, a phase modulating system wherein a phase of a carrier is changed in response to a modulating wave signal and an amplitude phase modulating system wherein an amplitude and a phase of a carrier are individually changed in response to a modulating wave signal.
The carrier signal (modulated signal) S(t), modulated in response to a modulating wave signal, generally expressed by the following equations. ##EQU1##
Herein, A(t) denotes an amplitude W.sub.c denotes a carrier frequency and .phi.(t) denotes a phase of a modulating wave signal.
As is clear from the above-described equation (1), the modulated signal can be represented by two components orthogonal to each other, that is, by a sum of an in-phase (I phase) component (the first term of the above-described equation (1)) and a quadrature phase (Q phase) component (the second term of the above-described equation (1)). Such a modulated signal can be therefore formed by using a quadrature modulator.
FIGS. 1 and 2 are a block diagram and a graph, respectively. These FIGURES, in combination, show the principle of such a quadrature modulator. It should be noted that the following example shows a phase modulating system for changing a phase of a carrier in response to a baseband signal, wherein an amplitude A (t) is fixed to "1".
With reference to FIG. 1, a mapping circuit 2 outputs I phase and Q phase components of a modulating wave signal as rectangular signals in response to a digital baseband signal applied through an input terminal 1. The I phase component is applied to one input of a multiplier 7 through a low pass filter (LPF) 3, while the Q phase component is applied to one input of a multiplier 8 through a low pass filter (LPF) 4.
A carrier signal cos.omega..sub.c t is applied from a signal source 5 to the other input of the multiplier 7 which outputs an I phase component sin.phi.(t).multidot.cos.omega..sub.c t of a modulated signal. A signal sin.omega..sub.c t obtained by shifting the phase of the carrier signal from the signal source 5 by .pi./2 by means of a phase shift circuit 6 is applied to the other input of the multiplier 8 which outputs an Q phase component cos.phi.(t).multidot.sin.omega..sub.c t of the modulated signal. The resulting I phase component and Q phase component can be represented in a one-to-one correspondence on the I and Q coordinates as shown in FIG. 2.
These I phase component and Q phase component are added to each other by an adder 9 to become such a modulated signal as expressed by equation (1), which signal is output from an output terminal 10.
FIG. 3 is a block diagram showing a GMSK (Gaussian filtered Minimum Shift Keying) modulator as an example of the quadrature modulator shown in FIG. 1. Such GMSK modulator is disclosed in, for example, "Differential Detection of GMSK Using Decision Feedback" by Abbas Yongacoglu et al., IEEE TRANSACTIONS ON COMMUNICATIONS, VOL. 36, No. 6, JUN. 1988, PP. 641-649 and "GMSK Modulation for Digital Mobile Radio Telephony" by Kazuaki Murota et al., IEEE TRANSACTIONS ON COMMUNICATIONS, VOL. COM-29, No.7, JULY 1981, pp. 1044-1050. In FIG. 3, the portion 2 surrounded by the chain dotted line shows the details of the structure corresponding to the mapping circuit 2 and LPFs 3 and 4 of FIG. 1.
First, a digital baseband signal applied through the input terminal 1 is applied to a gaussian filter 14. More specifically, the gaussian filter 14 comprise register 14a a binary counter 14b and a ROM-type low pass filter 14c. Shift register 14a takes the digital baseband signal R bits at a time and converts the R bits of digital baseband signal into R bit parallel data in response to a first clock signal applied from a clock signal source (not shown) through an input terminal 12 for every bit period T. The binary counter 14b counts and generates S bit data in response to a second clock signal applied from the clock, signal source (not shown) through an input terminal 13. The second clock signal has a frequency higher than that of the first clock signal. The ROM type low pass filter 14c from which is read L bit data indicating a phase shift amount .DELTA..phi.(t) of a modulating wave signal. The filter 14c is accessed by the R bit output data of the shift register 14a as a higher order address and the S bit output data of the binary counter 14b as a lower order address.
The L bit data indicating the phase shift amount .DELTA..phi.(t) is applied to an integration circuit 15 including an adder 15a and a one-clock delay unit 15b. The integration circuit 15 integrates the applied phase shift amount .DELTA..phi.(t), outputs P bit data indicating the phase .phi.(t) of the modulating wave and applies the data to ROMs 16 and 17.
The ROM 16 includes a ROM table including I phase component data of the modulating wave signal ROM 16 generates the corresponding I phase component data of W bits in response to the data indicating the phase .phi.(t) from the integration circuit 15 as an address. The ROM 17 includes a ROM table including Q phase component data of the modulating wave signal ROM 17 generates the corresponding Q phase component data of W bits in response to the data indicating the phase .phi.(t) from the integration circuit 15 as an address.
The digital I phase component data output from the ROM 16 is converted into an analog I phase component signal sin.phi.(t) by a D/A converter 18 and applied to one input of the multiplier 7 through the LPF 3'. The digital Q phase component data output from the ROM 17 is converted into an analog Q phase component signal cos.phi.(t) by a D/A converter 19 and applied to one input of the multiplier 8 through the LPF 4'. The subsequent operation is the same as previously described in connection with FIG. 1 and the terminal 10 outputs the modulated signal expressed by equation (1).
Such a conventional quadrature modulator as shown in FIG. 3 is structured such that the digital modulating wave component data (the outputs of the ROMs 16 and 17) are once converted into analog modulating component signals by the D/A converters 18 and 19, which are multiplied by carrier signals in an analog manner. In such a structure, if the signal gains of the I phase component and Q phase component are different from each other in the stages subsequent to the LPFs 3 and 4 in FIG. 3, the spatial coordinates of the signal points are not located on the unit circle of a radius of .sqroot.2 of FIG. 2 and the signal locus becomes an ellipse. In such a case, the accurate modulating wave signal components can not be obtained.
In addition, if the phases of two carrier components cos.omega.ct and sin.omega.ct whose phases are shifted by .pi./2 from each other are not precisely controlled, a satisfactory modulated signal can not be obtained.
In addition, since analog signal processing circuits are included, such circuit structure of the quadrature modulator as a whole can not be suitably made into an integrated circuit.